FM analog demodulator compatible with IBOC signals

ABSTRACT

A method includes: receiving an FM radio signal including an analog-modulated portion; digitally sampling an analog-modulated portion of the radio signal to produce a plurality of samples; using a ratio between an average magnitude and an RMS magnitude of a block of the samples to compute a signal quality metric; detecting sum and difference components of the baseband multiplex signal content; using the baseband content to produce an output signal; and blending the output signal from stereo to monaural as the signal quality metric falls below a threshold value.

FIELD OF THE INVENTION

This invention relates to radio broadcasting receivers, and moreparticularly to methods and apparatus for FM analog demodulation that iscompatible with in-band on-channel broadcasting signals.

BACKGROUND OF THE INVENTION

Digital radio broadcasting technology delivers digital audio and dataservices to mobile, portable, and fixed receivers. One type of digitalradio broadcasting, referred to as in-band on-channel (IBOC) digitalaudio broadcasting (DAB), uses terrestrial transmitters in the existingMedium Frequency (MF) and Very High Frequency (VHF) radio bands. HDRadio™ Technology, developed by iBiquity Digital Corporation, is oneexample of an IBOC implementation for digital radio broadcasting andreception.

IBOC DAB signals can be transmitted in a hybrid format including ananalog modulated carrier in combination with a plurality of digitallymodulated carriers, or in an all-digital format wherein the analogmodulated carrier is not used. Using the hybrid mode, broadcasters maycontinue to transmit analog AM and FM simultaneously with higher-qualityand more robust digital signals, allowing themselves and their listenersto convert from analog to digital radio while maintaining their currentfrequency allocations. IBOC DAB hybrid and all-digital waveforms aredescribed in U.S. Pat. No. 7,933,368, which is hereby incorporated byreference.

IBOC DAB technology can provide digital quality audio, superior toexisting analog broadcasting formats. Because each IBOC DAB signal istransmitted within the spectral mask of an existing AM or FM channelallocation, it requires no new spectral allocations. IBOC DAB promoteseconomy of spectrum while enabling broadcasters to supply digitalquality audio to the present base of listeners.

The National Radio Systems Committee, a standard-setting organizationsponsored by the National Association of Broadcasters and the ConsumerElectronics Association, adopted an IBOC standard, designated NRSC-5, inSeptember 2005. NRSC-5, the disclosure of which is incorporated hereinby reference, sets forth the requirements for broadcasting digital audioand ancillary data over AM and FM broadcast channels. The standard andits reference documents contain detailed explanations of theRF/transmission subsystem and the transport and service multiplexsubsystems. iBiquity's HD Radio Technology is an implementation of theNRSC-5 IBOC standard.

FM analog receivers implemented with digital signal processor (DSP)algorithms offer high performance, and are common for car radioreceivers. They generally offer low distortion, good stereo separation,and often high sensitivity and selectivity. However, IBOC signals maycreate some new digital interference conditions that were notanticipated by the designers of the FM analog demodulators. Althoughthis digital interference is generally limited to localized receptionareas, or occurs infrequently under special signal conditions, there maystill be concern about the impact on analog service. Furthermore, theFCC has authorized an increase in digital signal injection power frompresently 20 dB below the analog FM host, to 10 dB, potentiallyincreasing the interference.

It would be desirable to have an FM demodulator that is effective ineliminating the effects of interference from IBOC signals.

SUMMARY

In one embodiment, a method includes: receiving an FM radio signalincluding an analog-modulated portion; digitally sampling ananalog-modulated portion of the radio signal to produce a plurality ofsamples; using a ratio between an average magnitude and an RMS magnitudeof a block of the samples to compute a signal quality metric; detectingsum and difference components of the baseband multiplex signal content;using the baseband content to produce an output signal; and blending theoutput signal from stereo to monaural as the signal quality metric fallsbelow a threshold value.

In another embodiment, an apparatus includes an input for receiving anFM radio signal including an analog-modulated portion; and processingcircuitry for digitally sampling an analog-modulated portion of theradio signal to produce a plurality of samples, using a ratio between anaverage magnitude and an RMS magnitude of a block of the samples tocompute a signal quality metric, detecting sum and difference componentsof the baseband content, using the baseband content to produce an outputsignal, and blending the output signal from stereo to monaural as thesignal quality metric falls below a threshold value.

In another embodiment, a method includes: receiving an in-bandon-channel radio signal including an analog-modulated portion and adigitally modulated portion; sampling the radio signal to produce aplurality of successive complex signal samples; determining a phasedifference between successive ones of the complex signal samplesrepresenting the analog FM signal; and using the phase difference toobtain an FM baseband multiplex signal

In another embodiment, an apparatus includes: an input for receiving anin-band on-channel radio signal including an analog-modulated portionand a digitally modulated portion; and processing circuitry for samplingthe radio signal to produce a plurality of successive complex signalsamples, determining a phase difference between successive ones of thecomplex signal samples representing the analog FM signal, and using thephase difference to obtain an FM baseband multiplex signal

In another embodiment, a method includes: receiving a radio signal;sampling and filtering the radio signal to isolate the FM portion;demodulating the FM portion of the received signal; applying a slidingwindow to the first, second, and third samples; and adjusting the valueof the second sample if the product of the second sample times the sumof the first and third samples is less than a threshold value.

In another embodiment, an apparatus includes: an input for receiving aradio signal; and processing circuitry for sampling and filtering theradio signal to isolate the FM portion, demodulating the FM portion ofthe received signal, applying a sliding window to the first, second, andthird samples, and adjusting the value of the second sample if theproduct of the second sample times the sum of the first and thirdsamples is less than a threshold value.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic diagram of the FM baseband multiplex signal.

FIG. 2 is a simplified functional block diagram of a conventional FMreceiver with a stereo multiplex decoder.

FIG. 3 is a schematic representation of a hybrid FM IBOC waveform.

FIG. 4 is a schematic representation of another hybrid FM IBOC waveform.

FIG. 5 is a schematic representation of an all-digital FM IBOC waveform.

FIG. 6 shows spectral plots (one-sided) of an actual transmitted FMsignal and its input baseband multiplex signal.

FIG. 7 is a functional block diagram of isolation filters.

FIG. 8 is a halfquarter filter spectrum.

FIG. 9 shows the frequency response of an analog FM filter created fromthe cascade of halfquarter and halfband filters.

FIG. 10 is a decimated output of the analog FM filter showing passbandspectrum and aliasing due to decimation by 4.

FIG. 11 is an FM demodulator functional block diagram.

FIG. 12 is a comparison of the linear frequency response of an idealderivative function to the difference function for several sample rates.

FIG. 13 is a pilot phase locked loop (PLL) for recovery of a stereosubcarrier local oscillator (LO).

FIG. 14 is a parametric control signal functional block diagram.

FIG. 15 is a performance plot without difference compensation.

FIG. 16 is a performance plot with difference compensation.

FIG. 17 is an overmodulation performance plot without differencecompensation or click repair.

FIG. 18 is an overmodulation performance plot with differencecompensation and click repair.

DETAILED DESCRIPTION

In various aspects, the following description describes an FMdemodulator including a low sample rate, predetection filtering toaccommodate extended IBOC subcarriers, click and modulus overflowmitigation, blend-to-analog as a function of predetection signal quality(e.g., first-adjacent digital interference), and stereo-subcarriernonlinearity correction.

To understand the context of the invention, it is instructive to reviewa conventional analog FM signal and an FM receiver. Prior to frequencymodulation at the transmitter, an FM stereo multiplex signal isgenerated. A spectral representation of an FM baseband multiplex signal10 is presented in FIG. 1. The monophonic signal includes Left plusRight (L+R) components 12 (also called the sum component) that reside atbaseband from near zero up to 15 kHz. The stereo difference signalcomponent (L−R) is modulated (Double-Sideband Suppressed Carrier, DSBSC)at 38 kHz, forming an L−R lower sideband 14 and an L−R upper sideband16. A pilot signal 18 is placed at 19 kHz, coherent with half the 38 kHzsubcarrier. This aids in demodulation at the receiver. OptionalSubsidiary Communications Authorization (SCA) signals for auxiliaryservices 20 and 22 may also be included.

Notice that the stereo audio components of this signal extend to 53 kHz.The optional SCAs are also shown for informational purposes only, butmay be affected by the bandwidth reduction methods discussed later.

A simplified functional diagram of a conventional FM stereo receiver 30is shown in FIG. 2. A signal received on antenna 32 enters the low noiseamplifier (LNA) and preselection filter 34 to bring the signalsufficiently above the noise floor of the subsequent components, whileeliminating most unwanted signals. The local oscillator (LO) 36 andmixer 38 translate the signal frequency to a nominal IF (typically 10.7MHz) on line 40 for subsequent FM detection in the FM predetectionfilter/amplifier 42 and the FM detector 44 to output the FM basebandmultiplex signal on line 46. The 19-kHz pilot is recovered via aphase-locked loop (PLL) 48, where it is doubled in frequency to producethe coherent 38-kHz subcarrier on line 50. The 38-kHz subcarrier is usedto demodulate the L−R signal via multiplier 52. Next, the sum (L+R) anddifference (L−R) signal components are dematrixed using summation points54 and 56. The resulting Left and Right audio signals are thenbandlimited to 15 kHz and de-emphasized in blocks 58 and 60 to producethe Left and Right audio output signals.

It is important that the relative levels of the sum (L+R) and difference(L−R) signal components are accurately maintained to preserve stereoseparation. For example, if they are imbalanced by 1 percent, then thestereo separation is limited to 40 dB. The selectivity of the FMintermediate frequency (IF) predetection filter can be adjusted toaccommodate the interference conditions. Furthermore, the L−R signalinto the dematrix can be suppressed to further reduce the noise added bythe stereo L−R component. Well-designed receivers will adaptively adjustboth the predetection and postdetection bandwidths, and “blend out” theL−R signal to improve the compromise between postdetection noise andaudio fidelity in the presence of noise or interference.

IBOC DAB Waveforms

IBOC DAB signals can be transmitted in both AM and FM radio bands, usinga variety of waveforms. The waveforms include hybrid FM IBOC DABwaveforms and an FM all-digital IBOC DAB waveform.

FIG. 3 is a schematic representation of a hybrid FM IBOC waveform 70.The waveform includes an analog modulated signal 72 located in thecenter of a broadcast channel 74, a first plurality of evenly spacedorthogonally frequency division multiplexed subcarriers 76 in an uppersideband 78, and a second plurality of evenly spaced orthogonallyfrequency division multiplexed subcarriers 80 in a lower sideband 82.The digitally modulated subcarriers are divided into partitions andvarious subcarriers are designated as reference subcarriers. A frequencypartition is a group of 19 OFDM subcarriers containing 18 datasubcarriers and one reference subcarrier.

The hybrid waveform includes an analog FM-modulated signal, plusdigitally modulated primary main subcarriers. The subcarriers arelocated at evenly spaced frequency locations. The subcarrier locationsare numbered from −546 to +546. In the waveform of FIG. 3, thesubcarriers are at locations +356 to +546 and −356 to −546. Each primarymain sideband is comprised of ten frequency partitions. Subcarriers +546and −546, also included in the primary main sidebands, are additionalreference subcarriers. The amplitude of each subcarrier can be scaled byan amplitude scale factor.

FIG. 4 is a schematic representation of an extended hybrid FM IBOCwaveform 90. The extended hybrid waveform is created by adding primaryextended sidebands 92, 94 to the primary main sidebands present in thehybrid waveform of FIG. 3. One, two, or four frequency partitions can beadded to the inner edge of each primary main sideband. The extendedhybrid waveform includes the analog FM signal plus digitally modulatedprimary main subcarriers (subcarriers +356 to +546 and −356 to −546) andsome or all primary extended subcarriers (subcarriers +280 to +355 and−280 to −355).

The upper primary extended sidebands include subcarriers 337 through 355(one frequency partition), 318 through 355 (two frequency partitions),or 280 through 355 (four frequency partitions). The lower primaryextended sidebands include subcarriers −337 through −355 (one frequencypartition), −318 through −355 (two frequency partitions), or −280through −355 (four frequency partitions). The amplitude of eachsubcarrier can be scaled by an amplitude scale factor.

FIG. 5 is a schematic representation of an all-digital FM IBOC waveform100. The all-digital waveform is constructed by disabling the analogsignal, fully expanding the bandwidth of the primary digital sidebands102, 104, and adding lower-power secondary sidebands 106, 108 in thespectrum vacated by the analog signal. The all-digital waveform in theillustrated embodiment includes digitally modulated subcarriers atsubcarrier locations −546 to +546, without an analog FM signal.

In addition to the ten main frequency partitions, all four extendedfrequency partitions are present in each primary sideband of theall-digital waveform. Each secondary sideband also has ten secondarymain (SM) and four secondary extended (SX) frequency partitions. Unlikethe primary sidebands, however, the secondary main frequency partitionsare mapped nearer to the channel center with the extended frequencypartitions farther from the center.

Each secondary sideband also supports a small secondary protected (SP)region 110, 112 including 12 orthogonal frequency division multiplexing(OFDM) subcarriers and reference subcarriers 279 and −279. The sidebandsare referred to as “protected” because they are located in the area ofspectrum least likely to be affected by analog or digital interference.An additional reference subcarrier is placed at the center of thechannel (0). Frequency partition ordering of the SP region does notapply since the SP region does not contain frequency partitions.

Each secondary main sideband spans subcarriers 1 through 190 or −1through −190. The upper secondary extended sideband includes subcarriers191 through 266, and the upper secondary protected sideband includessubcarriers 267 through 278, plus additional reference subcarrier 279.The lower secondary extended sideband includes subcarriers −191 through−266, and the lower secondary protected sideband includes subcarriers−267 through −278, plus additional reference subcarrier −279. The totalfrequency span of the entire all-digital spectrum is 396,803 Hz. Theamplitude of each subcarrier can be scaled by an amplitude scale factor.The secondary sideband amplitude scale factors can be user selectable.Any one of four may be selected for application to the secondarysidebands.

In each of the waveforms, the digital signal is modulated usingorthogonal frequency division multiplexing (OFDM). OFDM is a parallelmodulation scheme in which the data stream modulates a large number oforthogonal subcarriers, which are transmitted simultaneously. OFDM isinherently flexible, readily allowing the mapping of logical channels todifferent groups of subcarriers.

In the hybrid waveform, the digital signal is transmitted in primarymain (PM) sidebands on either side of the analog FM signal in the hybridwaveform. The power level of each sideband is appreciably below thetotal power in the analog FM signal. The analog signal may be monophonicor stereo, and may include subsidiary communications authorization (SCA)channels.

In the extended hybrid waveform, the bandwidth of the hybrid sidebandscan be extended toward the analog FM signal to increase digitalcapacity. This additional spectrum, allocated to the inner edge of eachprimary main sideband, is termed the primary extended (PX) sideband.

In the all-digital waveform, the analog signal is removed and thebandwidth of the primary digital sidebands is fully extended as in theextended hybrid waveform. In addition, this waveform allows lower-powerdigital secondary sidebands to be transmitted in the spectrum vacated bythe analog FM signal.

Noise and Interference Issues

A brief summary of the noise and interference issues is described next.Noise can be characterized as additive white Gaussian noise (AWGN), suchas thermal noise at the front end of the receiver, or background noise.This noise is spectrally flat entering the predetection filter in thereceiver. FM detection of the baseband signal in the presence of AWGNresults in a non-flat output noise spectrum, where the postdetectionnoise density is proportional to the square of the frequency from zeroHz. For example, the postdetection noise density at 15 kHz is 225 times(23.5 dB greater than) the noise density at 1 kHz. This degradation ismitigated by de-emphasis of the output audio signal in the receiver formonophonic reception. Unfortunately, this de-emphasis is not aseffective for the stereo L−R signal centered at 38 kHz. The noisedensity centered at 38 kHz is a factor of 1444 (31.6 dB) greater thanthe noise density at 1 kHz, while the noise at 53 kHz is a factor of2809 (34.5 dB) greater. It can be shown that the noise contributed bythe upper sideband (USB) in the 38 to 53 kHz range is 3.4 dB greaterthan the lower sideband (LSB) noise from 23 to 38 kHz.

FM transmitter simulations have been performed with the L and R audiocomponents comprised of filtered Gaussian noise with 30% modulation,typical of audio processing in an FM system. The audio Gaussian noisewas filtered with a linear taper in frequency from 0 dB near 0 Hz to −12dB at 15 kHz, to simulate a typical audio spectral characteristic. Theinput audio stereo separation was set at 50%, meaning that 50% of thesignal was common in both the L and R components, while the remaining50% was uncorrelated.

FIG. 6 shows spectral plots (one-sided) of an actual transmitted FMsignal and its input baseband multiplex signal. FIG. 6 shows a one-sided(positive frequencies only) spectral representation illustrating thespectral compactness of the simulated baseband multiplex signal to 53kHz, while the frequency-modulated output extends well beyond that. Thespectral components of the FM output signal beyond 100 kHz are ofspecial interest here since these components affect the extended digitalsubcarriers in the extended hybrid modes of an IBOC DAB waveform.

Predetection Filter

Predetection filter characteristics play an important role in receiverperformance. The predetection filter affects stereo separation and audiodistortion. Since predetection filter bandwidth determines how muchnoise enters the FM detector, it also affects sensitivity. Perhaps mostimportantly, bandwidth control can reduce interference fromfirst-adjacent signals and the digital subcarriers of its own hybridIBOC sidebands. Since subcarriers in the extended hybrid mode can be asclose as ±101 kHz from FM center frequency, it is important that thefilter has adequate stopband attenuation beyond about 100 kHz. Thisfilter should also have linear phase (e.g., FIR) and be flat over thepassband up to about ±90 kHz, corresponding to 120% modulation. Typicalanalog ceramic IF filters have a significantly wider bandwidth thandigital filters, to minimize distortion and accommodate tolerances andgroup-delay variations due to their nonlinear phase characteristic.

Efficient Filter Design

Although primarily interested in FM predetection filter characteristics,a typical IBOC receiver design would also accommodate the digitalsidebands in an efficient architecture. Efficient implementation ofisolation filters, and decimation to minimum sample rates, can reducesubsequent processing requirements and save power. Recognition of somecomplementary characteristics of these filters is crucial to realizingthe efficient design opportunity. The fortuitous combination of inputsample rate, bandwidths, and locations of the analog FM signal anddigital sidebands, along with the decimate-by-4 frequencies, offers aconvenient filter architecture.

This unique set of characteristics allows exploitation of an inputfilter having both halfband and quarterband symmetries. FIG. 7 is afunctional block diagram of pre-detection isolation filters.

The “halfquarter” filter 120 establishes the locations of all thetransition bands between the passband and stopband of the filter. Thisis followed by an efficient halfband Hilbert-transform filter 122 toseparate the upper and lower digital sidebands, and another similarhalfband filter 124 to separate and reduce the sample rate of the analogFM signal. An upper sideband preacquisition filter 126 and a lowersideband preacquisition filter 128 are also included. In FIG. 7, allsignals are complex, and all filters are real, except the Hilbert FIRfilter.

Only those filters comprising the analog FM predetection filter, i.e.,the halfquarter and analog halfband FIR filters, are described in thefollowing sections.

The halfquarter filter efficiently establishes the transition bands forthe signal components. Its special symmetry results in nonzerocoefficients at every fourth filter coefficient, a very efficientstructure.

Integer versions of the filter coefficients are presented in Table 1,showing only one-sided coefficients starting at center index 0 through58. These integer coefficients would be multiplied by 2⁻¹⁵ for a unitypassband DC gain. The negative-indexed coefficients are equal to thepositive-indexed coefficients. Although this filter has 117coefficients, only 31 are nonzero. The symmetry of the upper and lowerhalves can be further exploited to halve the number of multiplies, afterfolding and adding the input signal samples. Then only 32 realmultiplies per sample are needed to filter the complex input signal.

TABLE 1 Positive-Indexed Coefficients of Halfquarter Filter Coefficients0 through 58 of Halfquarter Filter, Starting with Center Coefficient 016384 1 0 2 10396 3 0 4 0 5 0 6 −3374 7 0 8 0 9 0 10 1919 11 0 12 0 13 014 −1263 15 0 16 0 17 0 18 879 19 0 20 0 21 0 22 −624 23 0 24 0 25 0 26442 27 0 28 0 29 0 30 −309 31 0 32 0 33 0 34 210 35 0 36 0 37 0 38 −13839 0 40 0 41 0 42 85 43 0 44 0 45 0 46 −49 47 0 48 0 49 0 50 25 51 0 520 53 0 54 −11 55 0 56 0 57 0 58 4

The magnitude spectrum of the halfquarter filter and its complement(1-halfquarter) are shown in FIG. 8. The plot shows the response overthe Nyquist bandwidth for the complex sample rate fs=744.1875 kHz.Notice that the baseband passband spans the quarterband bandwidth ±fs/8,from −93 kHz to +93 kHz. This band carries the analog FM signal.

The pair of quarterband bandwidths carrying the digital sidebands iscreated from the complement of the halfquarter filter, from 93 kHz to279 kHz on either side of center frequency. These digital sidebandsaccommodate the extended subcarriers and additional bandwidth requiredfor a first-adjacent cancellation (FAC) operation. The three quarterbandpassbands, for analog FM and the pair of digital sidebands, are easilyand efficiently separated with subsequent filters, due to the largetransition bandwidths between passbands of the halfquarter filter.

Halfband Filter for Analog FM

The halfband filter efficiently captures the analog FM signal from theoutput passband of the halfquarter filter. The spectrum of this filterhas halfband symmetry, with alternating coefficients equal to zero.Integer versions of these filter coefficients are presented in Table 2,showing only one-sided coefficients starting at center coefficient index0 through 15. These integer coefficients would be multiplied by 2⁻¹⁵ fora unity passband gain. The negative-indexed coefficients are equal tothe positive-indexed coefficients.

TABLE 2 Positive-Indexed Coefficients of Halfband Analog FM FilterCoefficients 0 through 15 of Halfband Filter, Starting with CenterCoefficient 0 16384 1 10292 2 0 3 −3080 4 0 5 1479 6 0 7 −741 8 0 9 34310 0 11 −131 12 0 13 34 14 0 15 −4

The magnitude spectrum of the analog FM filter created from the cascadeof halfquarter and halfband filters is shown in FIG. 9. The plots showthe undecimated responses over the Nyquist bandwidth for the complexinput sample rate fs=744.1875 kHz, although only the decimated FM outputis computed (for efficiency). These plots include the output spectrum ofthe halfquarter filter, the spectrum of the halfband filter, and the FMfilter resulting from the cascade of the halfquarter and halfbandfilters. Notice that the baseband 6-dB passband spans the quarterbandbandwidth ±fs/8, from −93 kHz to +93 kHz. This band carries the analogFM signal.

The output sample rate of the halfband filter may be one-third orone-quarter of the input sample rate. Decimation by 4, for instance,results in very efficient filtering of the FM signal, but alsointroduces some aliasing, as shown in the magnitude spectrum of thehalfband filter in FIG. 10. In a separate analysis, outside the scope ofthis description, these aliasing effects were evaluated and shown to beacceptable for FM demodulation performance. Notice that the 6-dBfrequencies (±93 kHz) of the filter occur at ±½ the output sample rate,or the Nyquist bandwidth.

Decimation by 3 for this filter avoids aliasing at the extremes of thepassband created by the decimation by 4; however, the complex outputsample rate is 248.0625 ksps instead of 186.046875 ksps, increasing theMIPS requirement for subsequent FM demodulation.

FM Demodulator

The FM demodulator described herein is suitable for demodulating an FManalog signal (no digital components) and does not require that thereceived signal be an IBOC signal. However, it is also effective insuppressing the interference of a first-adjacent IBOC interferencesignal (its digital sideband).

A functional block diagram of the FM demodulator 140 is shown in FIG.11. The input 142 is the complex baseband signal from the FMpredetection filter, sampled at either approximately 186 kHz or 248 kHz.The outputs 144 and 146 include the left and right audio samples at 46.5kHz (or sample-rate-converted to 44.1 kHz), and an FM Analog SignalQuality Metric (ASQM) 148 for other uses such as additional audioprocessing or antenna diversity. The major components of the FMdemodulator include the FM detector 150, pilot PLL 152, stereodematrixing 154, de-emphasis filtering 156, 158, ASQM 160, andparametric control 162 for the blend-to-mono function.

Several uncommon features designed to enhance the performance of the FMdemodulator include compensation for the FM differential (instead ofderivative) detector, nonlinear compensation, “click repair,” pilotparametric control signals, and the use of ASQM for the blend-to-monometric. The latter feature is especially important to mitigatefirst-adjacent IBOC digital sideband interference in areas nearfirst-adjacent transmitters (spaced 200 kHz from the desired FM signal).

The advantages of a lower sample rate and predetection bandwidth includereduced computational load, lower predetection noise, immunity tofirst-adjacent interference, and elimination of digital sidebandinterference to its analog FM host signal. Thus, especially for hybridFM IBOC signals, it would be advantageous to use a lower sample rate(i.e., 186 kHz) and a lower predetection bandwidth. As described below,this FM demodulator addresses the following issues associated with lowersample rates:

-   -   1. Reduced stereo separation due to non-ideal approximation of        the derivative with a differential, and/or non-flat predetection        filtering.    -   2. Increased distortion (THD) associated with lower predetection        filter bandwidth.    -   3. Increased probability of modulo overflow with reduced sample        rate, resulting in a one-sample click.    -   4. Increased aliasing associated with reduced sample rate        (Nyquist bandwidth). While this problem could be eliminated by        increasing the sample rate (e.g., 248 kHz instead of 186 kHz),        that would increase processing requirements, and simulation        results indicate that the aliasing at 186 kHz is not a        significant problem.

FM Detection

FM detection is ideally accomplished by computing the continuousderivative of the phase of the predetection signal. Fordigitally-implemented receivers, FM detection can be approximated bycomputing the angle between successive complex signal samples, which isthe phase difference. It is convenient to first conjugate multiply eachpair of successive samples s_(n) and s_(n-1), then compute the angleover ±π of the result to get x_(n). The angle is the difference in phasebetween successive complex FM input samples. After proper scaling(described below), it represents the demodulated baseband multiplexsignal.

Appropriate scaling is applied to the resulting angle for the samplerate f_(s) and 100% deviation f_(d). Then the FM baseband multiplexsignal x_(n) has a range of ±1 with 100% modulation or frequencydeviation. This can be computed as:

Δ s_(n) = s_(n) ⋅ s_(n − 1)^(*)$x_{n} = \frac{a\;{{\tan\left( {\Delta\; s_{n}} \right)} \cdot {fs}}}{2 \cdot \pi \cdot f_{d}}$

Differential and Non-Linear Compensation

The approximation of the derivative of the phase by the phase differenceof consecutive samples results in an error seen in the non-flatfrequency response of the differentiator output. This error is increasedas the sample rate fs is reduced. The frequency response of thederivative function is:derivative(f)=2·π·f.

The frequency response of the difference function is:

${{diff}\left( {f,{fs}} \right)} = {{{fs} \cdot {{1 - {\mathbb{e}}^{{- j} \cdot 2 \cdot \pi \cdot {f/{fs}}}}}} = {2 \cdot {fs} \cdot {{\sin\left( \frac{\pi \cdot f}{fs} \right)}.}}}$

The plot of FIG. 12 compares the linear frequency response of an idealderivative function to the difference function over a range ofinstantaneous frequencies f (horizontal axis) for several sample rates.The sample rate fs=744.1875 kHz, while the decimated sample ratesinclude the lowest of fs/4=186.046875 kHz.

It is clear from the plots of FIG. 12 that the lower the sample rate,the more roll-off occurs at the higher frequencies. One straightforwardmethod of fixing this is to apply a compensation FIR filter tocounteract the gain droop at the higher frequencies in the FM basebandmultiplex signal; however, this may be costly in terms of processingrequirements.

Since the droop over the monophonic (L+R) bandwidth (from 0 to 15 kHz)appears negligible, it may not be necessary to compensate this portionof the signal. Subsequent measurements indicate that the Total HarmonicDistortion (THD) is very good without monophonic compensation, and itsdistortion is very small.

The stereo difference signal (L−R) is located in the bandwidth from 23to 53 kHz, where noticeable gain droop occurs. FM stereo receivers relyon accurate (relative) levels of the sum (L+R) and difference (L−R)signals for good stereo separation after matrix decoding (dematrixing).The strategy applied here is to compensate the gain loss in thedifference (L−R) signal path with a constant gain.

Although the slope of the gain over the 23 to 53 kHz region of the L−Rsignal is not linear, it can be approximated as a linear functionresulting in a flat passband in the DSBSC demodulated output signal. Anydeviations from linear result in variation of stereo separation acrossthe audio bandwidth. So the gain can be accurately compensated formaximum stereo separation at a single frequency (e.g., at 1 kHz oneither side of the 38 kHz subcarrier), assuming slight variations instereo separation are acceptable over the audio bandwidth. Since stereoseparation is measured at an audio frequency of 1 kHz (where humans aresensitive), it makes sense to compensate the gain for that frequency.Since the differential gain droop is nearly linear over a range of audiofrequencies (±1 kHz) about 38 kHz, the gain can be compensated at 38kHz. The linear droop compensation can be computed at various decimationfactors D=1, 2, 3, or 4. For example,

${{compdroop}\left( {f,{fs}} \right)} = \frac{\pi \cdot f}{{fs} \cdot {\sin\left( \frac{\pi \cdot f}{fs} \right)}}$compdroop(38000, fs) = 1.004 compdroop(38000, fs/2) = 1.017compdroop(38000, fs/3) = 1.04 compdroop(38000, fs/4) = 1.072.

Limiting the bandwidth of the predetection filter can also causenonlinear distortion in the difference (L−R) signal. Since an FM signalhas a theoretically infinite bandwidth, any bandlimiting results innonlinear distortion. Although a wideband predetection filter can beused, this should be avoided for reasons previously stated. Highfrequencies that fall out of passband occur primarily when theinstantaneous frequency deviation is high. This has the effect ofsoft-limiting the peaks of the difference (L−R) signal. The distortioncaused by the soft-limiting effects can be compensated by applying anonlinear correction to the difference signal. Simulation results haveverified that the distortion is approximately quadratic (relative to theinstantaneous amplitude of the difference signal), and can becompensated with a complementary quadratic function.

The flat gain droop, as well as the nonlinear distortion, can becompensated with the following function:x _(n) =x _(n)·(compdroop+compnonlin·x _(n) ²),where x_(n) is a sample of the difference (L−R) signal, and compdroopand compnonlin are constants determined by the sample rate. Thesecompensation coefficients may need further adjustment if differentfilters are used in place of the example design. No compensation isneeded at high sample rates, so compdroop=1 and compnonlin=0 in thatcase. For the rates in this receiver example:fs=744.1875 kHz;fs; x _(n) =x _(n)·(1.004+0.21·x _(n) ²)fs/2; x _(n) =x _(n)·(1.017+0.25·x _(n) ²,fs/3; x _(n) =x _(n)·(1.04+0.30·x _(n) ²)fs/4; x _(n) =x _(n)·(1.072+0.30·x _(n) ²)

Click Repair

Aliasing could occur in the FM detector due to overmodulation when thephase angle between samples produced by conjugate multiplying successivecomplex input samples is increased above +π or below −π radians. This isdue to modulo-2π overflow in the arctangent function. So as the anglereaches and exceeds π, the polarity reversal at the π boundary switchesto −π. This is similar to two's complement overflow/underflow. As theinstantaneous frequency deviation of the predetection signal approaches±f_(s)/2, the angle difference approaches ±π, and the potential formodulo overflow exists. Also, with high signal deviations approaching±π, noise can cause an overflow/underflow condition.

The maximum range of x_(n) corresponding to ±π is ±1.24 for a samplerate of 186 kHz, corresponding to ±124% modulation. Similarly, themaximum range is ±1.65 (±165% modulation) at the 248-kHz sample rate.Clearly, the probability of exceeding the boundary is reduced at highersample rates. So a “click repair” feature is more useful at the lowersample rate. It is also important to recognize that a lower sample ratecould increase distortion in the output audio signal due to the limitedNyquist bandwidth of the predetection filter (as addressed above).

An effective method to detect and correct the single-sample overflow orunderflow condition is described in the following operation following FMdetection:if (x _(n) +x _(n-2))·x _(n-1)≦−1, then let x _(n-1)=0.5·(x _(n) +x_(n-2)).

This operation observes a sliding “window” of 3 successive samplesx_(n-2), x_(n-1), and x_(n). If the polarity of the center samplex_(n-1), is reversed relative to the sum of the samples on either side,then the left-hand side of the expression is negative. If this negativevalue is small (e.g., <<1), then this would likely be a normal signalcondition at a low signal level, where the signal is crossing zero.However, if the left-hand side of the expression is less than −1, thenit is more likely that a large signal has crossed the modulo −2· (±π)boundary. This overflow/underflow condition would result in a largeaudio click in the output audio signal. In this case the center sampleis “repaired” by setting it equal to the average of the adjacentsamples.

The threshold value of −1 was verified empirically, although it could beadjusted as a function of the sample rate. Furthermore, the repairexpression averaging the samples on either side of the detected clicksample could be modified. For example, the adjacent sample with thelargest magnitude could replace the average function. The click repairfeature was created for the purpose of minimizing the sample rate lowerthan typically used in digitally-implemented FM receivers; it is notneeded if the sample rate is significantly greater than roughly 300ksps.

Pilot PLL

The Pilot PLL 152 for recovery of the stereo subcarrier local oscillator(LO) shown in FIG. 13 includes a phase-error detector (complexmultiplier) 180, a loop filter 182, and a numerically controlledoscillator (NCO) 184. A frequency doubler 186 is applied to the NCOoutput to regenerate the 38-kHz stereo-subcarrier local oscillator (LO).The input signal on line 188 is the sequence of complex basebandmultiplex samples. The outputs include the 38-kHz stereo-subcarrier LOand the pilot inphase signal for use in the Pilot Parametric Controlfunction. This Pilot PLL feature is somewhat common in FM receivers, sothe discussion here is limited.

Parametric Control Signals

The parametric control signals are used to improve audio SNR when thesignal is degraded due to noise, interference, or multipath fading. Theimprovement is realized by blending the stereo signal to monophonicthrough reduction or elimination of the difference signal in the stereodematrixing. This is accomplished through an SBC (Stereo Blend Control)signal which weights the L−R component from 0 (no stereo) to 1 (fullstereo separation). FIG. 14 is a functional block diagram of theparametric control signals. The SBC value is determined primarily by twoparametric signals: the availability of the pilot (Pcarrier) on line200, and the Analog Signal Quality Metric (ASQM) on line 202. These twoparametric signals are combined in the Stereo Metric block 204. Thestereo metric is filtered in block 206, and then the SBC on line 208 isformed as a function of both the filtered and unfilteredstereometricfast signal. The SBC function is restricted over a range ofzero to one, and it reacts to either the unfiltered stereometricfast orthe slower filtered stereometric signal, whichever is lower in valueafter appropriate scaling and offsets. This allows the SBC to reactquickly to a large abrupt signal-quality loss, but react more slowly toa gradual deterioration in signal quality.

The inphase pilot signal is filtered in block 210, followed by a powermeasurement in block 212, and then filtered again in block 214 toprovide a measure of the pilot power Pcarrier, having a normalized rangefrom zero to approximately 1. The ASQM measures the signal quality ofthe FM predetection signal, as described in the following section. Thestereometricfast signal is a function of the product of the ASQM andPcarrier signals:stereometricfast=2·Pcarrier·ASQM−1.

ASQM Computation

The ASQM function is an FM analog signal quality metric that can be usedfor the blend-to-mono function in FM receivers, as well as antennadiversity switching. The ASQM value is computed for blocks of samples.The recommended block size should span about 1 OFDM symbol. This blocksize is both convenient and practical. It is convenient since someimplementations already process signals framed at the symbol rate. Theblock size is large enough to get a reasonably accurate result, andsmall enough to accommodate flat fading over the time span.

The ASQM computation exploits the constant modulus property of an FMsignal where, in the absence of signal corruption, each sample has aconstant magnitude. Both noise and selective fading cause variations inFM sample energies over the symbol span of K samples. The ASQM can alsobe affected by the bandwidth of the FM preselection filter. The idealASQM is based on the ratio between the average (mean) magnitude and theRMS magnitude. This ratio is raised to a power p so that subsequentaveraging of ASQM values over time is not biased from a nominalthreshold of about 0.5. The greatest slope and an inflection point inthe ASQM versus the carrier-to-noise-ratio (C/No) characteristic occurat this threshold. This also provides convenient scaling, similar toother metrics used in the antenna diversity algorithm.

$\begin{matrix}{{ASQM\_ ideal} = \left( \frac{mean}{rms} \right)^{p}} \\{= \left( \frac{\frac{1}{K} \cdot {\sum\limits_{k = 0}^{K - 1}{x_{k}}}}{\sqrt{\frac{1}{K} \cdot {\sum\limits_{k = 0}^{K - 1}{x_{k}}^{2}}}} \right)^{p}} \\{= {\left( \frac{\sum\limits_{k = 0}^{K - 1}\sqrt{{{Re}\left\{ x_{k} \right\}^{2}} + {{Im}\left\{ x_{k} \right\}^{2}}}}{\sqrt{K \cdot {\sum\limits_{k = 0}^{K - 1}\left\lbrack {{{Re}\left\{ x_{k} \right\}^{2}} + {{Im}\left\{ x_{k} \right\}^{2}}} \right\rbrack}}} \right)^{p}.}}\end{matrix}$

Since the RMS value is the root-sum-square of the average (mean)magnitude and the standard deviation of the magnitude over the symboltime, then the average magnitude per sample is always less than or equalto its RMS value. This property results in an ASQM value between zeroand one. When ASQM=1, then there is no signal corruption, and themagnitude is constant. The minimum value of ASQM=K^(−p/2) occurs whenthere is only one nonzero sample. For convenience, the ideal ASQMcomputation is modified to avoid square roots in a more practical usage.An exponent value is chosen to accommodate the desired threshold ofabout 0.5. The modified practical ASQM result behaves similarly to theideal.

${ASQM} = {\left( \frac{\left( {\sum\limits_{k = 0}^{K - 1}\left\lbrack {{{Re}\left\{ x_{k} \right\}^{2}} + {{Im}\left\{ x_{k} \right\}^{2}}} \right\rbrack} \right)^{2}}{K \cdot {\sum\limits_{k = 0}^{K - 1}\left\lbrack {{{Re}\left\{ x_{k} \right\}^{2}} + {{Im}\left\{ x_{k} \right\}^{2}}} \right\rbrack^{2}}} \right)^{8}.}$

An ASQM value greater than about 0.5 generally indicates good signalquality, with maximum signal quality approaching 1. ASQM values lessthan 0.5 are indicative of poor signal quality, with the lowest qualityapproaching 0.

Improved ASQM Computation

The above ASQM computation is based on the ratio of the square of themean to the mean of the squared values of the signal magnitude-squared.However, since the magnitude is positive and cannot have a zero mean,then the ASQM cannot reach zero. An exponent power of 8 was used tosuppress smaller values of the ASQM. It can be shown that although theASQM approaches one for an ideal uncorrupted FM signal, noise only(AWGN) yields a value of one half to the exponent power of 8.

${{\begin{matrix}\lim \\\left. K\rightarrow\infty \right.\end{matrix}\frac{\left( {\sum\limits_{k = 0}^{K - 1}\left\lbrack {u^{2} + v^{2}} \right\rbrack} \right)^{2}}{K \cdot {\sum\limits_{k = 0}^{K - 1}\left\lbrack {u^{2} + v^{2}} \right\rbrack^{2}}}} = \frac{\left( {{E\left\{ u^{2} \right\}} + {E\left\{ v^{2} \right\}}} \right)^{2}}{{E\left\{ u^{4} \right\}} + {{2 \cdot E}\left\{ {u^{2} \cdot v^{2}} \right\}} + {E\left\{ v^{4} \right\}}}};$but  E{u²} = E{v²} = σ², and  E{u⁴} = E{v⁴} = 2 ⋅ σ⁴  (normal);${{Then}\mspace{14mu}\begin{matrix}\lim \\\left. K\rightarrow\infty \right.\end{matrix}\frac{\left( {\sum\limits_{k = 0}^{K - 1}\left\lbrack {u^{2} + v^{2}} \right\rbrack} \right)^{2}}{K \cdot {\sum\limits_{k = 0}^{K - 1}\left\lbrack {u^{2} + v^{2}} \right\rbrack^{2}}}} = {\frac{E\left\{ u^{2} \right\}^{2}}{E\left\{ u^{4} \right\}} = \frac{1}{2}}$

A simple adjustment to the previous ASQM expression extends the rangeover zero to one, with a target threshold of 0.5.

${ASQM} = \left( {\frac{2 \cdot \left( {\sum\limits_{k = 0}^{K - 1}\left\lbrack {{{Re}\left\{ x_{k} \right\}^{2}} + {{Im}\left\{ x_{k} \right\}^{2}}} \right\rbrack} \right)^{2}}{K \cdot {\sum\limits_{k = 0}^{K - 1}\left\lbrack {{{Re}\left\{ x_{k} \right\}^{2}} + {{Im}\left\{ x_{k} \right\}^{2}}} \right\rbrack^{2}}} - 1} \right)^{8}$

Additional details of the signal quality metric are shown in commonlyassigned U.S. patent application Ser. No. 13/165,325, filed Jun. 21,2011, for a “Method And Apparatus For Implementing Signal QualityMetrics And Antenna Diversity Switching Control”, which is herebyincorporated by reference.

Other techniques have been used to determine the FM analog signalquality for the purpose of blending to mono, but they have not beeneffective. Existing techniques for determining analog signal qualityinclude Received Signal Strength Indication (RSSI) and/or measuring thelevel of ultrasonic noise in the FM baseband multiplex signal,particularly close in frequency to the 19-kHz pilot. These existingmetrics are not sufficient to deal with the higher levels ofinterference local to a first-adjacent IBOC signal. Lab testing of a setof modern car receivers showed that these techniques do not force thereceiver to blend from stereo to a monaural signal at the expected RSSIlevel when the first-adjacent interference is large. The result isexcess audio noise received in the geographic vicinity of thefirst-adjacent IBOC interferer. The RSSI simply determines signal leveland assumes a noise floor (the noise floor is not necessarily measured).The ultrasonic noise is not a reliable indicator of interference noisesince stereo audio processing used at the transmitter site can placenoise artifacts in this ultrasonic region. Furthermore,frequency-selective fading of the received signal also places noise(distortion) in this (ultrasonic) region. However, the ASQM describedherein is able to reliably detect frequency-selective fading (multipathdistortion) or first-adjacent interference. The SBC is designed to blendappropriately under these conditions.

FM Receiver Performance

The plots in FIGS. 15-18 show various performance metrics of the analogFM demodulator. The predetection filter has a Nyquist bandwidth of 186kHz (±93 kHz, complex). This is established by the isolation filter,designed to effectively suppress any signals 100 kHz or greater from thecenter frequency. A 57 kHz unmodulated subcarrier, representative of theRBDS signal imposing a ±2 kHz deviation on the main carrier, wasincluded on all plots with 100% or greater modulation. A transmittedstereo signal is indicated when stereo=1 on the plot. AWGN was added tothe signals over a range of 50 to 100 dB_Hz carrier-to-noise-densityratio (C/No in units of dB_Hz), the horizontal axis variable. Ofparticular interest is the value of ASQM as a function of C/No for bothof these scenarios. Notice that the ASQM for stereo plots with 100%modulation (and greater) reach a limit noticeably less than 1. This isdue to the filtering of frequencies of the FM signal greater than thebandwidth of the predetection filter. FIG. 15 shows the performancewithout stereo difference compensation, while FIG. 16 shows performancewith stereo difference compensation and click repair, both at 100%modulation. Notice the large improvement in both stereo separation andsignal-to-noise and distortion ratio (SINAD) with the differencecompensation of FIG. 16.

FIGS. 17 and 18 show results at 120% modulation without and withdifference compensation and click repair, respectively. Notice that theclick repair is effective in extending the low range of CNR before FMthreshold, primarily for the 120% overmodulation case. The stereodifference compensation improves both SINAD in the stereo case, as wellas stereo separation. Also notice in this overmodulation case that theASQM is always less than 0.7. Filtering of the extended “overmodulated”bandwidth results in amplitude modulation, which reduces the ASQM value.Then this reduces the stereo separation, as appropriate for thisovermodulation case.

The various signal-processing methods described above can be implementedin a radio receiver or other apparatus having an input for receiving theradio signal and one or more processors or other processing circuitryfor performing the signal processing needed to implement the methods.

While the present invention has been described in terms of severalembodiments, it will be understood by those skilled in the art thatvarious modifications can be made to the disclosed embodiments withoutdeparting from the scope of the invention as set forth in the claims.

What is claimed is:
 1. A method comprising: receiving an in-bandon-channel radio signal including an analog-modulated portion and adigitally modulated portion; approximating a derivative of a phase ofthe radio signal by sampling the radio signal to produce a plurality ofsuccessive complex signal samples and determining a phase differencebetween successive ones of the complex signal samples representing theanalog FM signal; using the phase difference to obtain an FM basebandmultiplex signal, wherein approximating the derivative results in anerror in a non-flat frequency response of a differentiator output; andcompensating for gain loss in a stereo difference signal portion locatedin a bandwidth from 23 to 53 kHz of the FM baseband multiplex signalwith a constant gain to correct for the error resulting fromapproximating the derivative of the phase of the radio signal.
 2. Themethod of claim 1, further comprising: conjugate multiplying pairs ofsuccessive samples; and computing the angle over ±π of the result. 3.The method of claim 1, further comprising: scaling the phase differenceangle to be within a range of ±1 with 100% modulation or frequencydeviation to obtain the FM baseband multiplex signal.
 4. The method ofclaim 1, further comprising: decimating a rate of the complex signalsamples, wherein the compensation step applies a different gain fordifferent decimation factors.
 5. The method of claim 1, furthercomprising: filtering the in-band on-channel radio signal prior todetection of baseband content of the radio signal.
 6. A methodcomprising: receiving an in-band on-channel radio signal including ananalog-modulated portion and a digitally modulated portion; filteringthe in-band on-channel radio signal prior to detection of basebandcontent of the radio signal to produce a predetection signal; samplingthe predetection signal to produce a plurality of successive complexsignal samples; determining a phase difference between successive onesof the complex signal samples representing the analog FM signal; usingthe phase difference to obtain an FM baseband multiplex signal, whereinthe FM baseband multiplex signal is distorted due to limiting abandwidth of the predetection signal; and compensating a stereodifference signal portion located in a bandwidth from 23 to 53 kHz ofthe FM baseband multiplex signal to correct for distortion due tolimiting a bandwidth of the predetection signal, wherein non-linearityin the stereo difference signal portion is compensated with acomplementary quadratic function.
 7. The method of claim 6, wherein thefiltering is performed using a filter having a passband of about ±90 kHzand a stopband of about ±100 kHz.
 8. The method of claim 5, wherein thefiltered radio signal is sampled at approximately 186 kHz and thesampled signal is used to produce an output signal.
 9. An apparatuscomprising: an input for receiving an in-band on-channel radio signalincluding an analog-modulated portion and a digitally modulated portion;and processing circuitry for approximating a derivative of a phase ofthe radio signal by sampling the radio signal to produce a plurality ofsuccessive complex signal samples and determining a phase differencebetween successive ones of the complex signal samples representing theanalog FM signal, and wherein the processing circuitry uses the phasedifference to obtain an FM baseband multiplex signal, whereindetermining the phase difference results in an error in the non-flatfrequency response of a differentiator output; and compensates for gainloss in a stereo difference signal portion located in a bandwidth from23 to 53 kHz of the FM baseband multiplex signal with a constant gain tocorrect for an error resulting from approximating the derivative of thephase of the radio signal.
 10. The apparatus of claim 9, furthercomprising: conjugate multiplying pairs of successive samples; andcomputing the angle over ±ÿ of the result.
 11. The method of claim 9,further comprising: scaling the phase difference angle to be within arange of ±1 with 100% modulation or frequency deviation to obtain the FMbaseband multiplex signal.
 12. The apparatus of claim 9, wherein theprocessing circuitry decimates a rate of the complex signal samples, andapplies a different gain for different decimation factors.
 13. Theapparatus of claim 9, further comprising: filtering the in-bandon-channel radio signal prior to detection of baseband content of theradio signal.
 14. The apparatus of claim 13, wherein the filtering isperformed using a filter having a passband of about ±90 kHz and astopband of about ±100 kHz.
 15. The apparatus of claim 13, wherein thefiltered radio signal is sampled at approximately 186 kHz and thesampled signal is used to produce an output signal.
 16. An apparatuscomprising: an input for receiving an in-band on-channel radio signalincluding an analog-modulated portion and a digitally modulated portion;a filter for filtering the in-band on-channel radio signal prior todetection of baseband content of the radio signal to produce apredetection signal; and processing circuitry for sampling the radiosignal to produce a plurality of successive complex signal samples,determining a phase difference between successive ones of the complexsignal samples representing the analog FM signal, and using the phasedifference to obtain an FM baseband multiplex signal, wherein the FMbaseband multiplex signal is distorted due to limiting a bandwidth ofthe predetection signal, and wherein non-linearity in a stereodifference signal located in a bandwidth from 23 to 53 kHz of the FMbaseband multiplex signal is compensated with a complementary quadraticfunction to correct for distortion due to limiting a bandwidth of thepredetection signal.